The multicarrier modulation technique, associated for example with an error correction coding technique and interlacing, provides an efficient solution to the problem of broadcasting or transmission of information, for example in a radiomobile environment. Thus, the COFDM (Coded Orthogonal Frequency Division Multiplexing) modulation technique was selected for the DAB (Digital Audio Broadcasting), DVB-T (Digital Video Broadcasting—Terrestrial) and HIPERLAN/2 (High Performance Local Area Network) standards.
The multicarrier modulation used in COFDM system described for example in French patent document No. FR 2 765 757, comprises aparticularly simple equalization system based on the insertion of a guard interval. This guard interval, also called a cyclic prefix, behaves well faced with echoes, at the cost of a loss in spectral efficiency. New multicarrier modulations are currently being studied in order to avoid this loss, or at least to reduce it. Among these new multicarrier modulations, the invention is particularly applicable to OFDM/OQAM (Orthogonal Frequency Division Multiplexing/Offset Quadrature Amplitude Modulation) modulation, for which the carriers are shaped by Iota prototype function. Note that the Iota prototype function, described for example in patent document No. FR 2 733 869, has the characteristic that it is identical to its Fourier transform. The invention is obviously equally applicable to any other type of multicarrier modulation, particularly of the OFDM/OQAM type regardless of the associated prototype function.
Obviously, the process for shaping an electrical signal starting from the information to be transmitted depends on the conditions under which such a signal is transmitted. The characteristics of a transmission channel are summarised briefly below, particularly in a radiomobile environment, in order to better understand the advantage of using multicarrier modulations on such a channel.
In a radiomobile environment, the emitted wave is reflected many times along its route, and therefore the receiver receives a sum of delayed versions of the emitted signal. Each of these versions is attenuated and its phase is shifted at random. This phenomenon, known under the term “delay spread”, generates inter-symbol interference (ISI). For example, in an urban type environment, delay spreading is of the order of a few microseconds or less.
The receiver (for example the mobile radiotelephone of a car driver) is assumed to be moving, and hence the Doppler effect also acts on each path, which results in a frequency shift in the received spectrum proportional to the movement speed of the receiver. Note that there are also other types of Doppler effects, all of which can be taken into account by the technique implemented according to the invention.
A combination of these effects results in a non-stationary transmission channel with deep fading at some frequencies (therefore a frequency selective channel is obtained). For some applications, particularly useful for the purposes of the invention, the transmission band is wider than the channel coherence band (in other words the band for which the channel frequency response can be considered as being constant for a given duration). Therefore, fading appears in the band, in other words, at a given moment, some frequencies of the band are strongly attenuated.
In order to overcome these different phenomena (due to the ISI and the Doppler effect), it was envisaged to add a guard interval, particularly in OFDM type systems, during which information will not be transmitted so as to guarantee that all received information originates from the same symbol. In the case of a coherent demodulation of sub-carriers, the distortion due to the channel is then corrected by estimating its value at all points in the time-frequency network.
The introduction of such a guard interval reduces the problems related to inter-symbol interference, but one disadvantage of this technique according to prior art is that its spectral efficiency is low, since no useful information is transmitted during the guard interval.
Therefore, for the purposes of the invention, an attempt was made to find a technique capable of reducing the inter-symbol interference affecting multicarrier signals, without introducing a guard interval.
The main characteristics of a multicarrier modulation are summarised below, to give better understanding of interference phenomena between symbols and/or carriers of a multiplex. A multicarrier modulation is above all a digital modulation, in other words a process for generation of an electromagnetic signal, starting from digital information to be transmitted. The innovation and the advantage of such a modulation is to cut the frequency band allocated to the signal into a plurality of sub-bands, chosen with a width narrower than the channel coherence band, and on which the channel can therefore be considered as being constant during the transmission period of a symbol. Digital information to be transmitted during this period is then distributed on each of the sub-bands, so as to:                reduce the modulation speed (in other words increase the symbol duration) without modifying the transmitted throughput;        make a simple model of the action of the channel on each of the sub-bands, with reference to the complex multiplier model.        
In reception, an uncomplicated system for correction of received data (consisting of making a complex division by the estimated channel) makes it possible to satisfactorily recover information emitted on each of the carriers, except for carriers for which deep fading occurred. In this case, if no measures are taken to protect the information, the data transported by these carriers will be lost. Therefore, a multicarrier system is only interesting if generation of the electrical signal is preceded by digital data processing, for example such as error correction coding and/or interlacing.
In particular, at the moment, two types of orthogonal multicarrier modulations are known, for example described in patent document No. FR 2 733 869, and for which the characteristics are summarised below.
All carriers in a multicarrier modulation form a multiplex. Each of the carriers of this multiplex is shaped using the same prototype function denoted g(t) that characterises the multicarrier modulation. We will use ν0 to denote the spacing between two adjacent multiplex carriers, and τ0 to denote the time spacing between two emitted multicarrier symbols. The signal emitted at each instant nτ0 on the mth sub-band with central frequency νm is am,neiφm,ne2iτνmtg(t−nτ0), in which the am,n terms represent the digital data to be transmitted. The expression of the signal emitted in low band (centred around frequency Mν0) is then:
                              s          ⁡                      (            t            )                          =                              ∑            n                    ⁢                                    ∑                              m                =                0                                                              2                  ⁢                  M                                -                1                                      ⁢                                          a                                  m                  ,                  n                                            ⁢                              ⅇ                                                      i                    ⁢                                                                                  ⁢                                          φ                                              m                        ,                        n                                                                              ⁢                                                                                                    ⁢                              ⅇ                                  2                  ⁢                  i                  ⁢                                                                          ⁢                  π                  ⁢                                                                          ⁢                                      mv                    0                                    ⁢                  t                                            ⁢                              g                ⁡                                  (                                      t                    -                                          n                      ⁢                                                                                          ⁢                                              τ                        0                                                                              )                                                                                        (        I        )            
Note that we have considered the case of a signal with an even number of frequency sub-bands, for simplification purposes. Obviously, the signal could be written more generally in the following form:
      s    ⁡          (      t      )        =            ∑      n        ⁢                  ∑                  m          =          0                          M          -          1                    ⁢                        a                      m            ,            n                          ⁢                  ⅇ                                    i              ⁢                                                          ⁢                              φ                                  m                  ,                  n                                                      ⁢                                                                ⁢                  ⅇ                      2            ⁢            i            ⁢                                                  ⁢            π            ⁢                                                  ⁢                          mv              0                        ⁢            t                          ⁢                  g          ⁡                      (                          t              -                              n                ⁢                                                                  ⁢                                  τ                  0                                                      )                              
Remember that according to one conventional technique, digital data am,n with a zero value on the edges of the spectrum are input, which modifies the number of terms actually used in the above sum, and for example leads to an even number of carriers.
The gm,n(t)=eiφm,ne2iπmν0tg(t−nτ0)) functions are called “time-frequency” translated functions of g(t). To find the information transmitted by each sub-carrier, you need to choose g(t) and the phases φm,n such that the “time-frequency” translated functions given above can be separated. A sufficient condition to satisfy this separability property is that the translated functions should be orthogonal, in the sense of a scalar product defined on a set of finite energy functions (which is a Hilbert space in the mathematical sense).
Note that the following two scalar products can be calculated on the finite energy functions space:                the complex scalar product        
      〈          x      /      y        〉    =            ∫      R        ⁢                  x        ⁡                  (          t          )                    ⁢                        y          *                ⁡                  (          t          )                    ⁢              ⅆ                                  ⁢        t                            the real scalar product        
            〈              x        /        y            〉        R    =      ℜe    ⁢                  ∫        R            ⁢                        x          ⁡                      (            t            )                          ⁢                              y            *                    ⁡                      (            t            )                          ⁢                  ⅆ                                          ⁢          t                    
Thus, two types of multicarrier modulations are defined:                a complex type multicarrier modulation, for which the g(t) function chosen guarantees orthogonality of its translated functions in the complex sense. For example, this is the case for OFDM modulation, also called OFDM/QAM (Orthogonal Frequency Division Multiplexing/Quadrature Amplitude Modulation). For this type of modulation, φm,n=0 and the data am,n are complex.        a real time multicarrier modulation, for which the g(t) function chosen enables orthogonality of its translated functions in the real sense. For example, this is the case of the OFDM/OQAM, OFDM/OMSK (Offset Minimum Shift Keying) or OFDM/OQAM/IOTA modulations. For this type of modulation, φm,n=(π/2)*(m+n) and the am,n data are real.        
The characteristics of these two types of modulations induce significant differences, particularly in terms of the time-frequency network density associated with the modulation considered.
Remember that these multicarrier modulations are designed to transmit information at high speed, and consequently their spectral efficiency is fairly high, and in particular can reach 4 bits/Hz (for example in the case of a digital television). The transformation of bits output from an error correction encoder in modulation symbols (mapping) will thus be of the QAM (Quadrature Amplitude Modulation) type.
A complex variable output from the QAM constellation is therefore transmitted differently depending on the multicarrier modulation type used.
Thus, for a complex type modulation, the real and imaginary parts of a complex output from the QAM constellation are transmitted simultaneously, every symbol time Ts; on the other hand, in the case of a real type modulation, the real and imaginary parts are transmitted with a time offset of a half symbol time (Ts/2) (this is called the Offset QAM or OQAM).
Therefore, for a particular transmission band and for a given number of sub-carriers, the transmission rate of real type multicarrier symbols to carry the same throughput has to be twice the transmission rate of complex type multicarrier symbols.
Furthermore, these two information transmission modes are characterised by the density of the associated time-frequency network d=1/(ν0 τ0). Thus, real type multicarrier modulations correspond to a density d=2, whereas complex type multicarrier modulations correspond to a density d=1.
The distinct characteristics of firstly real type multicarrier modulations, and secondly complex type multicarrier modulations, require different processing when making an estimate of the transmission channel. In the case of a real type multicarrier modulation, and as described in the rest of this document, the channel estimating process is made more difficult because all that is available is an orthogonality of the translated functions in the real sense. In order to get a better understanding of this problem, we will now try to describe known channel estimating techniques for a multicarrier modulation as presented above.
In the rest of the reasoning, it is assumed that the multicarrier modulation parameters are chosen such that the channel can be considered as being practically constant on each of the sub-carriers (multiplication channel), for each OFDM symbol. The channel can then be modelled by a complex coefficient to be estimated, Hm,n (where m is the sub-carrier index and n is the index of the OFDM symbol considered).
One conventional technique for estimating the channel in OFDM consists of inserting reference carriers in the useful carrier flow, at positions known to the receiver. In reception, the values adopted by these reference carriers, called pilots, are read and it is easy to deduce the complex channel gain at these reference positions from them. The complex gain of the channel on all points in the transmitted time-frequency network is then derived from the calculated value of the complex gain at the reference positions.
In particular, in the context of the OFDM/QAM, a method was considered based on the use of an estimate by scattered pilots. The pilots are scattered according to a regular pattern in the time-frequency plane, and are used to measure an under sampled version of the channel. The next step is to make a two-dimensional interpolation to determine the value of the channel at all points in the time-frequency network. For example, this method is used in the DVB-T standard (Digital Video Broadcasting—DVB); Framing Structure, channel coding and modulation for digital terrestrial television (DVB-T), Digital Video Broadcasting (DVB); ETS 300 744, March 1997), and is illustrated in FIG. 1 in which 6 OFDM symbols numbered from 0 to 5 are shown. Each cross (x) represents a reference carrier, and each dot (.) represents useful data to be transmitted.
Therefore, the invention presented in this document is more particularly applicable to this method, called the scattered pilots channel estimating method.
In the case of an OFDM/OQAM (Offset QAM) type multicarrier modulation, the channel estimating process is made more difficult because all that is available is orthogonality of the translated functions in the real sense. To estimate the complex gain of the channel on a given sub-carrier, the complex projection of the signal received on the considered sub-carrier needs to be made. In this case, orthogonality of the translated functions in the real sense and the fact that the prototype functions, even if they are optimally located in time and frequency, extend to infinity over at least one of the time and frequency axes, means that there will be (intrinsic) interference between carriers, even on an ideal channel.
Within the framework of a real type multicarrier modulation, the imaginary part of the projection of the received signal based on translated functions of the prototype function is not zero. A disturbing term then appears, which is added to the demodulated signal, which will have to be corrected before the channel is estimated. Therefore, it is necessary to think of methods of compensating for this lack of complex orthogonality, thus overcoming the disadvantages of this technique according to prior art.
According to the technique presented above, the complex projection of a multicarrier signal r(t) received at point (m0, n0) in the time-frequency space is used to estimate the channel Ĥm0,n0 at this position. Thus, if √{square root over (E)} is emitted in (m0,n0), we have
            H      ^                      m        0            ,              n        0              =            ∫                        r          ⁡                      (            t            )                          ⁢                              g                                          m                0                            ,                              n                0                                      *                    ⁡                      (            t            )                          ⁢                  ⅆ                                          ⁢          t                            E      
Therefore, assuming that the channel is ideal (r(t)=S(t)), we should have Ĥm0,n0=1.
Now
                              ∫                                    s              ⁡                              (                t                )                                      ⁢                                          g                                                      m                    0                                    ,                                      n                    0                                                  *                            ⁡                              (                t                )                                                    =                              E                    +                                                    ∑                                                      (                                          m                      ,                      n                                        )                                    ≠                                      (                                                                  m                        0                                            ,                                              n                        0                                                              )                                                              ⁢                                                a                                      m                    ,                    n                                                  ⁢                                  ∫                                                                                    g                                                  m                          ,                          n                                                                    ⁡                                              (                        t                        )                                                              ⁢                                                                  g                                                                              m                            0                                                    ,                                                      n                            0                                                                          *                                            ⁡                                              (                        t                        )                                                                                                                                ︸                                                1                                                            m                      0                                        ,                                          n                      0                                                                      ⁢                                  ∈                  i                                ⁢                R                                                                        (        II        )            
Equation (II) represents the fact that the complex projection of the perfectly transmitted signal is nevertheless affected by ISI (Inter-symbol interference) intrinsic to OFDM/OQAM modulations. ISI includes interference between time symbols and/or between carriers.
The existence of this intrinsic ISI that disturbs the transmission channel estimate, is a major disadvantage of this technique according to prior art.
In particular, the purpose of the invention is to overcome these disadvantages according to prior art.
More precisely, one purpose of the invention is to supply a multicarrier modulation technique to reduce intrinsic interference between symbols and/or between carriers.
Another purpose of the invention is to implement a multicarrier modulation technique that is simple and costs little to implement.
Another purpose of the invention is to supply a multicarrier modulation technique adapted to OFDM/OQAM type systems.
Another purpose of the invention is to implement a multicarrier modulation technique through which the scattered pilot channel estimating method can be adapted to OFDM/OQAM type signals.
Another purpose of the invention is to supply a multicarrier modulation technique for implementing a scattered pilot channel estimating method that is more precise than methods used according to prior art.
Another purpose of the invention is to implement a multicarrier modulation technique enabling improved reception, demodulation and decoding of the emitted multicarrier signal.